Complex digital phase locked loop for use in a demodulator and method of optimal coefficient selection

ABSTRACT

A complex digital phase locked loop for use in a digital demodulator includes a phase detector for producing a phase error indicative of a difference in phase between a complex digital input signal and a complex digital feedback signal. The phase error is input to a controller, which multiplies the phase error by a gain factor selected to stabilize and optimize the phase locked loop and produces an output signal for use in extracting a frequency deviation present in the complex digital input signal. The output signal is also input to a numerically controlled oscillator that tracks the phase of the complex digital input signal based on the output signal and produces the complex digital feedback signal.

CROSS REFERENCE TO RELATED APPLICATIONS

The present U.S. Utility Patent Application claims priority pursuant to35 U.S.C. §120, as a continuation, to the following U.S. Utility PatentApplication, which is hereby incorporated herein by reference in itsentirety and made part of the present U.S. Utility Patent Applicationfor all purposes:

1. U.S. Utility application Ser. No. 10/999,531, entitled “COMPLEXDIGITAL PHASE LOCKED LOOP FOR USE IN A DEMODULATOR AND METHOD OF OPTIMALCOEFFICIENT SELECTION,”, filed Nov. 30, 2004, pending.

BACKGROUND

1. Technical Field

The present invention relates to wireless communications and, moreparticularly, wideband wireless communication systems.

2. Related Art

Communication systems are known to support wireless and wire linedcommunications between wireless and/or wire lined communication devices.Such communication systems range from national and/or internationalcellular telephone systems to the Internet to point-to-point in-homewireless networks. Each type of communication system is constructed, andhence operates, in accordance with one or more communication standards.For instance, wireless communication systems may operate in accordancewith one or more standards, including, but not limited to, IEEE 802.11,Bluetooth, advanced mobile phone services (AMPS), digital AMPS, globalsystem for mobile communications (GSM), code division multiple access(CDMA), local multi-point distribution systems (LMDS),multi-channel-multi-point distribution systems (MMDS), and/or variationsthereof.

Depending on the type of wireless communication system, a wirelesscommunication device, such as a cellular telephone, two-way radio,personal digital assistant (PDA), personal computer (PC), laptopcomputer, home entertainment equipment, etc., communicates directly orindirectly with other wireless communication devices. For directcommunications (also known as point-to-point communications), theparticipating wireless communication devices tune their receivers andtransmitters to the same channel or channels (e.g., one of a pluralityof radio frequency (RF) carriers of the wireless communication system)and communicate over that channel(s). For indirect wirelesscommunications, each wireless communication device communicates directlywith an associated base station (e.g., for cellular services) and/or anassociated access point (e.g., for an in-home or in-building wirelessnetwork) via an assigned channel. To complete a communication connectionbetween the wireless communication devices, the associated base stationsand/or associated access points communicate with each other directly,via a system controller, via a public switch telephone network (PSTN),via the Internet, and/or via some other wide area network.

Each wireless communication device includes a built-in radio transceiver(i.e., receiver and transmitter) or is coupled to an associated radiotransceiver (e.g., a station for in-home and/or in-building wirelesscommunication networks, RF modem, etc.). As is known, the transmitterincludes a data modulation stage, one or more intermediate frequencystages, and a power amplifier stage. The data modulation stage convertsraw data into baseband signals in accordance with the particularwireless communication standard. The one or more intermediate frequencystages mix the baseband signals with one or more local oscillations toproduce RF signals. The power amplifier stage amplifies the RF signalsprior to transmission via an antenna.

As is also known, the receiver is coupled to the antenna and includes alow noise amplifier, one or more intermediate frequency stages, afiltering stage, and a data recovery stage. The low noise amplifierreceives an inbound RF signal via the antenna and amplifies it. The oneor more intermediate frequency stages mix the amplified RF signal withone or more local oscillations to convert the amplified RF signal into abaseband signal or an intermediate frequency (IF) signal. As usedherein, the term “low IF” refers to both baseband and intermediatefrequency signals. A filtering stage filters the low IF signals toattenuate unwanted out of band signals to produce a filtered signal. Thedata recovery stage demodulates the filtered signal to recover the rawdata in accordance with the particular wireless communication standard.Alternate designs being pursued at this time further include directconversion radios that produce a direct frequency conversion often in aplurality of mixing steps or stages.

As an additional aspect, these designs are being pursued as a part of adrive to continually reduce circuit size and power consumption. Alongthese lines, such designs are being pursued with CMOS technology therebypresenting problems not addressed by prior art designs. For example, onecommon design goal is to provide an entire system on a single chip. Thedrive towards systems-on-chip solutions for wireless applicationscontinues to replace traditionally analog signal processing tasks withdigital processing to exploit the continued shrinkage of digital CMOStechnology.

One approach of current designs by the applicant and assignee herein isto reduce analog signal processing performance requirements and tocompensate for the relaxed performance requirements in the digitaldomain to provide required system performance. This approach isbeneficial in that, in addition to the reduced silicon arearequirements, the processing is insensitive to process and temperaturevariations.

Applications for which this trend is observed include RF receivers wherethe received signal is digitized as early as possible in the receiverchain using a high dynamic range analog-to-digital converter (ADC), andin a variety of calibration circuits of the radio where signal levelsmust be measured accurately over a wide range of values. This trend thusincreases the demand for embedded low-power, low-voltage ADCs providinghigh dynamic range in the interface between the analog and digitalprocessing. A class of ADCs capable of providing high dynamic range andparticularly suitable for low-power and low-voltage implementation isknown as continuous-time delta sigma analog-to-digital converters(CTΔΣADCs). These ADCs can be designed to operate with supply voltagesin the range 1.2V-1.5V and current consumption as low as a few hundredμAs.

With the introduction of CTΔΣADCs, digital demodulators are beingintroduced in the receiver architecture to replace traditional analogdemodulators. Digital processing, unlike analog processing, does notintroduce DC offset into the signal. As a result, the performance ofdigital demodulators is typically superior to that of analogdemodulators. However, the design of digital demodulators is specific tothe communication standard being employed. Therefore, an appropriatedemodulator design must be developed for each type of communicationstandard.

For example, the most widespread standard used in wireless personal areanetwork (PAN) communication is currently Bluetooth 1.1. This standardemploys the Gaussian Frequency Shift Keying (GMSK) modulation scheme,which is a constant-envelope binary modulation scheme, with a maximumraw transmission rate of 1 Megabits per second (Mpbs). Bluetooth furtheremploys a frequency hopping scheme for the purposes of sharing thespectrum resources and increasing the robustness towards undesiredinterference. Bluetooth devices operate in the 2.4 GHz unlicensedindustrial, scientific and medical (ISM) band and occupy an RF channelbandwidth of 1 MHz. However, an optimal digital demodulator design hasyet to be developed for the Bluetooth standard.

Thus, a need exists for a digital demodulator design for application ina Bluetooth receiver.

SUMMARY OF THE INVENTION

The present invention is directed to apparatus and methods of operationthat are further described in the following Brief Description of theDrawings, the Detailed Description of the Invention, and the claims.Other features and advantages of the present invention will becomeapparent from the following detailed description of the invention madewith reference to the accompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

A better understanding of the present invention can be obtained when thefollowing detailed description of the preferred embodiment is consideredwith the following drawings, in which:

FIG. 1 is a functional block diagram illustrating a communication systemthat includes a plurality of base stations or access points (APs), aplurality of wireless communication devices and a network hardwarecomponent;

FIG. 2 is a schematic block diagram illustrating a digital demodulatorfor use in a receiver according to one embodiment of the presentinvention;

FIG. 3 is a schematic block diagram illustrating a complex digital phaselocked loop for use in the digital demodulator according to oneembodiment of the present invention;

FIG. 4 is a circuit schematic illustrating an exemplary complex digitalphase locked loop for use in the digital demodulator according to oneembodiment of the present invention;

FIG. 5 illustrates a linearized model of the complex digital phaselocked loop according to one embodiment of the present invention;

FIG. 6 is a graph illustrating an exemplary stability region of thecomplex digital phase locked loop according to one embodiment of thepresent invention;

FIG. 7 is a flowchart illustrating one method of the present invention;

FIG. 8 is a graph illustrating exemplary discrete points of thestability region of FIG. 6 according to one embodiment of the presentinvention;

FIG. 9 is a graphical representation of exemplary bit error rate valuesfor the discrete points in the stability region of FIG. 8 with a lowsignal to noise (SNR) ratio according to one embodiment of the presentinvention; and

FIG. 10 is a graphical representation of exemplary bit error rate valuesfor the discrete points in the stability region of FIG. 8 with adjacentchannel interference according to one embodiment of the presentinvention.

DETAILED DESCRIPTION OF THE DRAWINGS

FIG. 1 is a functional block diagram illustrating a communication system10 that includes a plurality of base stations or access points (APs)12-16, a plurality of wireless communication devices 18-32 and a networkhardware component 34. The wireless communication devices 18-32 may belaptop computers 18 and 26, personal digital assistants 20 and 30,personal computers 24 and 32 and/or cellular telephones 22 and 28. Thedetails of the wireless communication devices will be described ingreater detail with reference to FIGS. 2-9.

The base stations or APs 12-16 are operably coupled to the networkhardware component 34 via local area network (LAN) connections 36, 38and 40. The network hardware component 34, which may be a router,switch, bridge, modem, system controller, etc., provides a wide areanetwork connection 42 for the communication system 10. Each of the basestations or access points 12-16 has an associated antenna or antennaarray to communicate with the wireless communication devices in itsarea. Typically, the wireless communication devices 18-32 register withthe particular base station or access points 12-16 to receive servicesfrom the communication system 10. For direct connections (i.e.,point-to-point communications), wireless communication devicescommunicate directly via an allocated channel.

Typically, base stations are used for cellular telephone systems andlike-type systems, while access points are used for in-home orin-building wireless networks. For example, access points are typicallyused in Bluetooth systems. Regardless of the particular type ofcommunication system, each wireless communication device and each of thebase stations or access points includes a built-in radio and/or iscoupled to a radio. The radio includes a transceiver (transmitter andreceiver) for modulating/demodulating information (data or speech) bitsinto a format that comports with the type of communication system.

FIG. 2 illustrates a high-level architecture of an exemplary digitaldemodulator 200 for use in a Bluetooth receiver. The digital demodulator200 includes a direct digital frequency synthesizer (DDFS) 210, a pairof digital multipliers 220 and 225, an in-phase low pass filter (I-LPF)230, a quadrature-phase low pass filter (Q-LPF) 235, a complex digitalphase locked loop (CDPLL) 240, an equalizer (LPF/Equalizer) 250 and adata and timing recovery circuit 260.

The digital demodulator is connected to receive a complex digitalmodulated signal 205. The complex digital modulated signal 205 includesan in-phase signal 208 and a quadrature-phase signal 209. In the exampleshown in FIG. 2, the input to the digital demodulator 200 is a 2-bit, 24MHz digitized version of the hard-limited intermediate frequency (IF)signal centered at 2 MHz. The DDFS 210 produces outputs that are used bythe pair of digital multipliers 200 and 225 to translate the complexdigital modulated IF signal to a complex digital modulated basebandsignal. The DDFS 210 outputs a digital cosine function 212 and a digitalsine function 214. The first digital multiplier 220 multiplies thein-phase signal 208 (at IF) by the digital cosine function 212 toproduce a first digital signal 222 (at baseband). The second digitalmultiplier 225 multiplies the quadrature-phase signal 209 (at IF) by thedigital sine function 214 to produce a second digital signal 228 (atbaseband). However, it should be understood that in other embodiments,the DDFS 210 may not be used or included if the input to the digitaldemodulator 200 is a complex digital modulated baseband signal. Inaddition, other IF translation circuitry may be included if the input tothe digital demodulator 200 is a complex digital modulated radiofrequency (RF) signal and an additional downconversion from RF to IF isrequired.

The I-LPF 230 is connected to receive the first digital signal 222 andis operable to filter the first digital signal 222 to produce a firstfiltered digital signal 232. The Q-LPF 235 is connected to receive thesecond digital signal 228 and is operable to filter the second digitalsignal 228 to produce a second filtered digital signal 238. For example,in one embodiment, the I-LPF 230 and Q-LPF 235 attenuate interferers andquantization noise from the in-phase and quadrature-phase modulatedbaseband signals 222 and 228, respectively, to produce the first andsecond filtered digital signals 232 and 238, respectively. In theexemplary embodiment shown in FIG. 2, the I-LPF 230 and Q-LPF 235 aredecimation filters that operate to both perform low pass filtering andreduce the digital sampling rate. For example, the decimation filtersmay lower the sampling rate from 24 MHz (the sampling rate of the inputto the digital demodulator 200) to 12 MHz (the sampling rate of theCDPLL 240).

The CDPLL 240 is connected to receive the first and second filtereddigital signals 232 and 238, respectively, and operates to demodulatethe first and second filtered digital signals 232 and 238 to produce anoutput signal 240 that is used by the data and timing recovery circuit260 to extract the frequency deviations present in the complex digitalinput signal 205. The frequency deviations correspond to the modulationused in the transmitter (e.g., Gaussian Frequency Shift Keying “GFSK”).The output signal 245 of the CDPLL 240 is input to the LPF/Equalizer250, which is operable to attenuate the noise component of the CDPLLoutput 245 and boost the gain of the parts of the signal that sufferfrom filtering in the transmitter and receiver. Thus, the LPF/Equalizer250 operates as a smoothing filter that applies a smoothing function tothe output 245 of the CDPLL 240 to produce a smoothed output signal 255.

In one embodiment, the LPF/Equalizer 250 is a decimation filter thatoperates to both perform filtering and reduce the digital sampling rate.For example, as shown in FIG. 2, the LFP/Equalizer 250 lowers thesampling rate from 12 MHz (the sampling rate of the output of the CDPLL240) to 6 MHz (the sampling rate of the data and timing recovery circuit260). The smoothed output signal 255 is input to the data and timingrecovery circuit 260, which is operable to determine the optimalsampling points of the smoothed output signal 255 and to produce ademodulated digital signal 270. The demodulated digital signal 270includes digital baseband bits representative of the originaltransmitted digital data.

FIG. 3 illustrates a top-level block diagram of a 2^(nd) order CDPLL 240appropriate for application in GFSK demodulation, according to oneembodiment of the invention. The CDPLL 240 includes a multiplier 310, aphase detector 320, a proportionality/integration (PI)-controller 330and a numerically controlled oscillator (NCO) 340. The digital inputsignal, s[n], is assumed to be of the complex form:s[n]=e ^(j(ω) ^(LO) ^(n+θ) ^(s) ^([n])),where ω_(LO) denotes an arbitrary center frequency and θ_(s)[n] denotesthe signal phase deviation.

The NCO 340 generates a complex digital feedback signal of the form:t[n]=e ^(−j(ω) ^(LO) ^(n+θ) ^(NCO) ^([n])),where θ_(NCO)[n] denotes the NCO phase deviation. The digital multiplier310 multiples the complex digital input signal s[n] and the complexdigital feedback signal t[n] to produce a multiplied digital signal v[n]of the form:v[n]=e ^(j(θ) ^(s) ^([n]−θ) ^(NCO) ^([n])),The multiplied digital signal v[n] is input to the phase detector 320,which is operable to determine a phase error signal w[n] indicative ofthe difference in phase between the complex digital input signal s[n]and the complex digital feedback signal t[n]. The phase error signalw[n] has the form:w[n]=sin(θ_(s) [n]−θ _(NCO) [n]).The phase error signal w[n] is input to the PI-controller 330, which isoperable to multiply the phase error signal w[n] by a gain factorselected to stabilize and optimize the complex digital phase locked loop240. The output of the PI-controller 330, y[n], is input to the NCO 340to produce the complex digital feedback signal t[n] based on the outputsignal y[n]. The closed loop action of the loop causes the phase errorsignal w[n] to approach zero. Hence, the phase of the output signal y[n]tracks the phase of the complex digital baseband signal s[n], asdesired.

As will be shown in more detail below in connection with FIGS. 6-9, abroad set of gain parameters for the PI-controller 330 exist such thatthe steady-state response to a frequency-step input is proportional tothe magnitude of the frequency-step, and for a more restricted set ofgain parameters, the CDPLL 240 achieves error-free tracking in merelythree samples. To achieve these tracking properties, the purpose of thePI-controller 330 is to direct the NCO phase θ_(NCO)[n] to equal thesignal phase θ_(s)[n] in some optimal manner determined by the gainparameter. When this “phase-lock” occurs,θ_(s) [n]≈θ _(NCO) [n],and thusw[n]≈θ _(s) [n]−θ _(NCO) [n]≡θ _(c) [n],where θ_(e)[n] is the phase error signal w[n].

FIG. 4 is a circuit schematic illustrating an exemplary complex digitalphase locked loop 240 for use in the digital demodulator according toone embodiment of the present invention. The input signal s[n] and NCOoutput t[n] are in complex form. The phase detector 320 includes themultiplier 310 shown in FIG. 3, and therefore, includes a first digitalmultiplier 410, a second digital multiplier 420 and a subtractor 430.The first digital multiplier 410 is connected to receive an in-phaseinput signal 400 (corresponding to the first filtered digital signal 232in FIG. 2) and an in-phase feedback signal 490, and is operable tomultiply the in-phase input signal 400 with the in-phase feedback signal490 to produce a first digital signal 415. The second digital multiplier420 is connected to receive a quadrature-phase input signal 405(corresponding to the second filtered digital signal 238 in FIG. 2) anda quadrature-phase feedback signal 495, and is operable to multiply thequadrature-phase input signal 405 with the quadrature-phase feedbacksignal 495 to produce a second digital signal 425. The subtraction node430 is connected to receive the first digital signal 415 and the seconddigital signal 425, and is operable to subtract the first digital signal415 from the second digital signal 425 to produce the phase error signalw[n].

In the example shown in FIG. 4, the PI-controller 330 includes a firstpath 445 connected to receive the phase error w[n] and a second path 440connected to receive the phase error w[n]. The first path 445 includesan integration gain factor K_(I) selected to stabilize and optimize theCDPLL 240 and an integrator 450 operable to integrate a product 449 ofthe phase error w[n] and the integration gain factor K_(I) to produce anintegrated signal 455. The second path 440 includes a proportionalitygain factor K_(P) selected to stabilize and optimize the CDPLL 240 andan adder 460 operable to add a product 448 of the phase error w[n] andthe proportionality gain factor K_(P) and the integrated signal 455 toproduce the output signal y[n].

It should be understood that in other embodiments, the PI-controller 330may include no paths, only the first path 445, only the second path 440,or one or more additional paths with one or more correspondingadditional gain factors selected to stabilize and optimize the CDPLL240. For example, with a simple choice of gain-parameters, e.g.,(K_(I),K_(P),K_(NCO))=(1.00,1.00,1.00), the CDPLL 240 can be implementedwithout gain-factors. However, in practice, due to implementationrestrictions, the assumption of unity signal amplitude and unity NCOgain made in the above may not hold true and the PI-controller gainsmust be adjusted accordingly. In particular, let K_(S) denote the inputsignal amplitude, i.e.,s[n]=K _(S) e ^(j(ω) ^(LO) ^(n+θ) ^(s) ^([n])),and, as before, let K_(NCO) be a implicit gain associated with the NCOimplementation. To design for CDPLL deadbeat response, the PI-controllergains must satisfy:

$K_{I} = {K_{P} = {\frac{1}{K_{S}K_{NCO}}.}}$In general, to implement a second-order CDPLL 240 with PI-controllergains (K_(I),K_(P)), input signal amplitude K_(S), and implicit NCO gainK_(NCO) requires implementation of gains as follows:

$\frac{K_{P}}{K_{S}K_{NCO}},{and}$ $\frac{K_{I}}{K_{S}K_{NCO}}.$

The NCO 340 includes an integrator 470, an implicit gain K_(NCO) and asine/cosine look-up table (LUT) 480. The integrator 470 is connected toreceive the output signal y[n] of the PI-controller 330 and anadditional signal ω_(LO) that operates to remove intermediatefrequencies from the output signal y[nh]. If the complex digital inputsignal s[n] is at baseband, the additional signal ω_(LO) is set to zero.The integrator 470 produces an integrated signal 475, which ismultiplied by the intrinsic gain K_(NCO) of the NCO 340 and input to thesine/cosine LUT 480. The LUT 480 is operable to compute the complexdigital feedback signal t[n] based on a product 478 of the integratedsignal 475 and the intrinsic gain K_(NCO) of the NCO 340.

In phase-lock, the CDPLL 240 can be represented in linearized form asshown in FIG. 5. The term q[n] denotes additive quantization noiseresulting from finite-precision representation in the LUT, and, asdescribed above, K_(NCO) denotes a possible implicit gain associatedwith the implementation of the LUT. From FIG. 5, it can be seen that thenoise transfer function H(z) of the CDPLL 240 is of the form:

$\begin{matrix}{{H(z)} \equiv \frac{Y(z)}{\Theta_{s}(z)}} \\{= \frac{\left\lbrack {{K_{P}\left( {1 - z^{- 1}} \right)} + K_{I}} \right\rbrack\left( {1 - z^{- 1}} \right)}{\left( {1 - z^{- 1}} \right)^{2} + {\left\lbrack {{C_{P}\left( {1 - z^{- 1}} \right)} + C_{I}} \right\rbrack z^{- 1}}}} \\{{= \frac{K_{P} + K_{I} - {\left( {{2K_{P}} + K_{I}} \right)z^{- 1}} + {K_{P}z^{- 2}}}{1 + {\left( {C_{P} + C_{I} - 2} \right)z^{- 1}} + {\left( {1 - C_{P}} \right)z^{- 2}}}},}\end{matrix}$where the proportionality gain coefficient C_(P)=K_(P)×K_(NCO) and theintegration gain coefficient C_(i)=K_(I)×K_(NCO). The noise transferfunction H(z) determines the amount of attenuation of the in-bandportion of the “noise” terms θ_(e)[n] and q[n]. Thus, the stability ofthe CDPLL 240 depends upon both C_(P) and C_(I).

FIG. 6 is a graph illustrating an exemplary stability region 600 of thecomplex digital phase locked loop according to one embodiment of thepresent invention. For a stability region in R²(C_(I),C_(P)) for adiscrete-time system with characteristic polynomialP(z)=z ²+(C _(P) +C _(I)−2)z+(1−C _(P)),the system poles are given by:

${z = \frac{2 - C_{P} - {C_{I} \pm \sqrt{D}}}{2}},$where:D=(C _(P) +C _(I)−2)²−4(1−C _(P)).

For real-valued poles:D≧0

(C _(P) +C _(I)−2)²≧4(1−C _(P))

(C _(P) +C _(I))²≧4C _(I)This is trivially true for negative C_(I) and any C_(P). However, ifC_(I)≧0, then(C _(P) +C _(I))²≧2² C _(I)

C _(P) +C _(I)≧2√{square root over (C_(I))} or C _(P) +C _(I)≦−2√{squareroot over (C_(I))}.Thus, real-valued poles are achieved in the regionR _(REAL)(C _(I) ,C _(P))={C _(I) <C _(P) arbitrary}∪{C _(I)≧0,C_(P)≧2√{square root over (C _(I))}−C _(I) }∪{C _(I)≧0,C _(P)≦−2√{squareroot over (C _(I))}−C _(I)}and complex poles are achieved in the regionR _(COMPLEX)(C _(I) ,C _(P))=R ²(C _(I) ,C _(P))−R _(REAL)(C _(I) ,C_(P)).

For stable, real-valued poles,

${D \geq 0},{{{and}\mspace{14mu}{z}} = {{\frac{2 - C_{P} - {C_{I} \pm \sqrt{D}}}{2}} < 1.}}$

Thus, the values of C_(I) and C_(P) must satisfy:

$\underset{\underset{{Case}\; 1}{︸}}{{- 2} < {2 - C_{P} - C_{I} + \sqrt{D}} < 2}\mspace{14mu}{and}\mspace{14mu}{\underset{\underset{{Case}\; 2}{︸}}{{- 2} < {2 - C_{P} - C_{I} - \sqrt{D}} < 2}.}$Case 1 holds true for:

$\underset{\underset{{Case}\; 1.1}{︸}}{{2 - C_{P} - C_{I} + \sqrt{D}} < 2}\mspace{14mu}{and}\mspace{14mu}{\underset{\underset{{Case}\; 1.2}{︸}}{{- 2} < {2 - C_{P} - C_{I} + \sqrt{D}}}.}$For case 1.1:√{square root over (D)}<C _(P) +C _(I)

(C _(P) +C _(I)−2)²<(C _(P) +C _(I))²

C_(I)>0.For case 1.2:C _(P) +C _(I)−4<√{square root over (D)}.This is trivially satisfied for:C _(P) +C _(I)−4<0=

C _(I)<4−C _(P).Therefore, assuming that:C _(P) +C _(I)−4≧0,then:(C _(P) +C _(I)−4)²<(C _(P) +C _(I)−2)²−4(1−C _(P))

C _(I)>4−2C _(P).

Case 2 holds true for:

$\underset{\underset{{Case}\; 2.1}{︸}}{{2 - C_{P} - C_{I} - \sqrt{D}} < 2}\mspace{14mu}{and}\mspace{14mu}{\underset{\underset{{Case}\; 2.2}{︸}}{{- 2} < {2 - C_{P} - C_{I} - \sqrt{D}}}.}$For case 2.1:−C _(P) −C _(I) <√{square root over (D)}.This is trivially satisfied for:−C _(P) −C _(I)<0

C _(I) >−C _(P).Therefore, assuming that:−C _(P) −C _(I)≧0,then:(C _(P) +C _(I))²<(C _(P) +C _(I)−2)²−4(1−C _(P))

C_(I)>0.For case 2.2:√{square root over (D)}<4−C _(P) −C _(I)

(C _(P) +C _(I)−2)²−4(1−C _(P))<(C _(P) +C _(I)−4)²

C _(I)<4−2C _(P).

Combining the in-equalities established in Cases 1.1, 1.2, 2.1, and 2.2for the region R_(REAL)(C_(I),C_(P)) results in the stable, real-poleregion 610 indicated in FIG. 6. For stable, complex-valued poles,

${D < 0},{{{and}\mspace{14mu}{z}} = {{\frac{2 - C_{P} - {C_{I} \pm {j\sqrt{- D}}}}{2}} < 1.}}$${Thus},\left. {{{2 - C_{P} - C_{I} + {j\sqrt{- D}}}}^{2} < 4}\Rightarrow{{\left( {C_{P} + C_{I} - 2} \right)^{2} + {4\left( {1 - C_{P}} \right)} - \left( {C_{P} + C_{I} - 2} \right)^{2}} < 4}\Rightarrow{C_{P} > 0.} \right.$Applying the above in-equality in the region R_(COMPLEX)(C_(I),C_(P))results in the stable, complex-pole region 620 indicated in FIG. 6.

Therefore, the stability region 600, R_(STABLE)(C_(I),C_(P)), equals thetriangular region:

${{R_{STABLE}\left( {C_{I},C_{P}} \right)} = \left\{ {{0 < C_{I} < 4};{0 < C_{P} < {\frac{C_{I}}{2} + 2}}} \right\}},$depicted in FIG. 6.

From the stability region 600, R_(STABLE)(C_(I),C_(P)), it is difficultto arrive at an analytical, closed-form expression for the choice ofoptimal values of K_(I) and K_(P) for a number of reasons. First, manyperformance specifications exist for Bluetooth, and a determination ofpriority must be made among these specifications. Second, once apriority has been determined, the specific demodulator performancedepends upon a manifold of variables in the receive path: analogIF-filter, I/Q LPFs, LPF/Equalizer, choice of sample rates, etc.Therefore, to reduce problem complexity, in accordance with embodimentsof the invention, choices for the CDPLL gain parameters are madea-priori, and then the optimal CDPLL parameters are selected from thesechoices.

FIG. 7 is a flowchart illustrating one method of the present inventionfor selecting an optimal integration gain coefficient value C_(I) and anoptimal proportionality gain coefficient value C_(P) of the CDPLL.Initially, a stability region in the complex plane associated with theCDPLL is determined as a function of integration gain coefficient valuesand proportionality gain coefficient values (step 700), such as thestability region shown in FIG. 6. Thereafter, the stability region isdivided into discrete points, each representing a specific one of theintegration gain coefficient values and a corresponding specific one ofthe proportionality gain coefficient values (step 710).

An example of discrete points is shown in FIG. 8. FIG. 8 is a graphillustrating an exemplary optimization region of the stability region ofFIG. 6 according to one embodiment of the present invention. Theoptimization region includes a plurality of discrete points,corresponding to a gain parameter pair (C_(I), C_(P)). Referring againto FIG. 7, for each of the discrete points, the demodulator performanceis measured under strenuous operating conditions (step 720), and theoptimal the optimal integration gain coefficient value and the optimalproportionality gain coefficient value are chosen as the discrete pointfor which the best overall performance is observed (step 730).

Among the most important performance metrics are sensitivity andadjacent channel interference (ACI) rejection. In a simulation, for eachdiscrete point 800 in the stability region 600 shown in FIG. 8, a totalof 100,000 random bits were demodulated for the cases of low SNR (14.5dB IF SNR) and high ACI (−3.5 dB). FIG. 9 is a graphical representationof exemplary bit error rate values for the discrete points 800 in thestability region 600 of FIG. 8 with a low signal to noise (SNR) ratioaccording to one embodiment of the present invention. FIG. 10 is agraphical representation of exemplary bit error rate values for thediscrete points 800 in the stability region 600 of FIG. 8 with adjacentchannel interference according to one embodiment of the presentinvention.

For these simulations, the transmitted signal was Gaussian FSKcorresponding to a 520 kHz 4^(th) order Gaussian filter, a modulationindex of 0.5 and a frequency offset of 30 kHz. FIG. 9 and FIG. 10 showBER vs. value of CDPLL gain parameters (C_(I),C_(P)) for 14.5 dB IF SNRand −3.5 dB ACI, respectively. In both cases, K_(NCO)=1. The minimum BERfor 14.5 dB IF SNR of 0.08% is achieved for (C_(I),C_(P))=(0.75,1.50),while a minimum BER for −3.5 dB ACI of 0.05% is achieved for(C_(I),C_(P))=(0.25,1.75). Therefore, based on the results of thesesimulations, a near-optimal choice of gain parameters for the Bluetoothdemodulator would be (C_(I),C_(P))=(0.25,1.75).

While the invention is susceptible to various modifications andalternative forms, specific embodiments thereof have been shown by wayof example in the drawings and detailed description. It should beunderstood, however, that the drawings and detailed description theretoare not intended to limit the invention to the particular formdisclosed, but, on the contrary, the invention is to cover allmodifications, equivalents and alternatives falling within the spiritand scope of the present invention as defined by the claims. As may beseen, the described embodiments may be modified in many different wayswithout departing from the scope or teachings of the invention.

1. A complex digital phase locked loop for use in a demodulator,comprising: a phase detector connected to receive a complex digitalinput signal and a complex digital feedback signal, wherein the phasedetector is operable to produce a phase error indicative of a differencein phase between the complex digital input signal and the complexdigital feedback signal; a controller connected to receive the phaseerror, wherein the controller is operable to multiply the phase error bya gain factor selected to stabilize and optimize the complex digitalphase locked loop and produce an output signal for use in extracting afrequency deviation present in the complex digital input signal byanother component of the demodulator; and a numerically controlledoscillator connected to receive the output signal, wherein thenumerically controlled oscillator is operable to track the phase of thecomplex digital input signal based on the output signal and produce thecomplex digital feedback signal; wherein the gain factor is selectedfrom a set of gain parameters such that the steady-state response of thephase locked loop to a frequency-step change to the complex digitalinput signal is proportional to the magnitude of the frequency-step. 2.The phase locked loop of claim 1, wherein the gain factor is anintegration gain factor, and wherein the controller further includes anintegrator operable to integrate a product of the phase error and theintegration gain factor to produce an integrated signal.
 3. The phaselocked loop of claim 2, wherein the controller further includes a firstpath connected to receive the phase error, the first path including theintegration gain factor and the integrator, and a second path connectedto receive the phase error, the second path including a proportionalitygain factor selected to stabilize and optimize the phase locked loop andan additive node operable to add a product of the phase error and theproportionality gain factor and the integrated signal to produce theoutput signal.
 4. The phase locked loop of claim 3, wherein theintegration gain factor and the proportionality gain factor are selectedfrom a discrete set of respective values corresponding to a stabilityregion in the complex plane associated with the phase locked loop. 5.The phase locked loop of claim 1, wherein the numerically controlledoscillator includes an integrator connected to receive the output signaland an additional signal to remove intermediate frequencies from theoutput signal and produce an integrated signal, and wherein thenumerically controlled oscillator further includes a look-up table foruse in computing the complex digital feedback signal based on theintegrated signal and an intrinsic gain of the numerically controlledoscillator.
 6. The phase locked loop of claim 1, wherein the complexdigital input signal includes an in-phase input signal and aquadrature-phase input signal and the complex digital feedback signalincludes an in-phase feedback signal and a quadrature-phase feedbacksignal, and wherein the phase detector includes: a first digitalmultiplier connected to receive the in-phase input signal and thein-phase feedback signal, wherein the first digital multiplier isoperable to multiply the in-phase input signal with the in-phasefeedback signal to produce a first digital signal, a second digitalmultiplier connected to receive the quadrature-phase input signal andthe quadrature-phase feedback signal, wherein the second digitalmultiplier is operable to multiply the quadrature-phase input signalwith the quadrature-phase feedback signal to produce a second digitalsignal, and a subtraction node connected to receive the first digitalsignal and the second digital signal, wherein the subtraction node isoperable to subtract the first digital signal from the second digitalsignal to produce the phase error.
 7. The phase locked loop of claim 1,further comprising: a digital multiplier connected to receive thecomplex digital input signal and the complex digital feedback signal andoperable to multiply the complex digital input signal with the complexdigital feedback signal to produce a multiplied digital signal that isinput to the phase detector.
 8. The phase locked loop of claim 1,wherein the complex digital input signal is one of a complex digitalradio frequency (RF) input signal, a complex digital IntermediateFrequency (IF) input signal or a complex digital baseband input signal.9. A method for operating a complex digital phase locked loop within adigital demodulator, comprising: receiving a complex digital inputsignal and a complex digital feedback signal; producing a phase errorindicative of a difference in phase between the complex digital inputsignal and the complex digital feedback signal; selecting a gain factorfrom a set of gain parameters such that the steady-state response of thecomplex digital phase locked loop to a frequency-step change to thecomplex digital input signal is proportional to the magnitude of thefrequency-step; multiplying the phase error by the gain factor selectedto stabilize and optimize the complex digital phase locked loop toproduce a multiplied signal for use in extracting a frequency deviationpresent in the complex digital input signal by another component of thedemodulator; producing an output signal based on the multiplied signal,the output signal tracking the phase of the complex digital inputsignal; and producing the complex digital feedback signal from theoutput signal.
 10. The method of claim 9, wherein the gain factor is anintegration gain factor, and further comprising: integrating a productof the phase error and the integration gain factor to produce anintegrated signal.
 11. The method of claim 10, further comprising:adding a product of the phase error and a proportionality gain factorselected to stabilize and optimize the phase locked loop with theintegrated signal to produce the output signal.
 12. The method of claim11, further comprising: selecting the integration gain factor and theproportionality gain factor from a discrete set of respective valuescorresponding to a stability region in the complex plane associated withthe phase locked loop.
 13. The method of claim 9, further comprising:integrating the multiplied signal and an additional signal selected toremove intermediate frequencies from the output signal to produce anintegrated signal; and accessing a look-up table to compute the complexdigital feedback signal based on the integrated signal.
 14. The methodof claim 9, wherein the complex digital input signal includes anin-phase input signal and a quadrature-phase input signal and thecomplex digital feedback signal includes an in-phase feedback signal anda quadrature-phase feedback signal, and further comprising: multiplyingthe in-phase input signal with the in-phase feedback signal to produce afirst digital signal, multiplying the quadrature-phase input signal withthe quadrature-phase feedback signal to produce a second digital signal,and subtracting the first digital signal from the second digital signalto produce the phase error.
 15. The method of claim 9, wherein thecomplex digital input signal is one of a complex digital radio frequency(RF) input signal, a complex digital Intermediate Frequency (IF) inputsignal or a complex digital baseband input signal.